An electric power steering apparatus which provides a steering mechanism of a vehicle with a steering assist torque (an assist torque) by means of a rotational torque of a motor, applies a driving force of the motor as the assist torque to a steering shaft or a rack shaft by means of a transmission mechanism such as gears or a belt through a reduction mechanism. In order to accurately generate the steering assist torque, such a conventional electric power steering apparatus (EPS) performs a feedback control of a motor current. The feedback control adjusts a voltage supplied to the motor so that a difference between a steering assist command value (a current command value) and a detected motor current value becomes small, and the adjustment of the voltage applied to the motor is generally performed by an adjustment of duty command values of a PWM control.
A general configuration of a conventional electric power steering apparatus will be described with reference to FIG. 1. As shown in FIG. 1, a column shaft (a steering shaft, a handle shaft) 2 connected to a steering wheel (a steering handle) 1, is connected to steered wheels 8L and 8R through reduction gears 3, universal joints 4a and 4b, a rack and pinion mechanism 5, and tie rods 6a and 6b, further via hub units 7a and 7b. Further, the column shaft 2 is provided with a torque sensor 10 for detecting a steering torque of the steering wheel 1, and a motor 20 for assisting the steering force of the steering wheel 1 is connected to the column shaft 2 through the reduction gears 3. Electric power is supplied to a control unit (ECU) 100 for controlling the electric power steering apparatus from a battery 13, and an ignition key signal is inputted into the control unit 100 through an ignition key 11. The control unit 100 calculates a current command value of an assist (steering assist) command based on a steering torque Tr detected by the torque sensor 10 and a vehicle velocity Vel detected by a vehicle velocity sensor 12, and controls a current supplied to the motor 20 based on a voltage control value E obtained by performing compensation and so on with respect to the current command value. Moreover, it is also possible to receive the vehicle velocity Vel from a CAN (Controller Area Network) and so on.
The control unit 100 mainly comprises a CPU (or an MPU or an MCU), and general functions performed by programs within the CPU are shown in FIG. 2.
Functions and operations of the control unit 100 will be described with reference to FIG. 2. As shown in FIG. 2, the steering torque Tr detected by the torque sensor 10 and the vehicle velocity Vel from the vehicle velocity sensor 12 are inputted into a current command value calculating section 101, and a current command value Iref1 is calculated by means of an assist map and so on. The calculated current command value Iref1 is inputted into a maximum output limiting section 102 and an output is limited based on an overheat protection condition or the like in the maximum output limiting section 102. A current command value Iref2 that a maximum output is limited, is inputted into a subtracting section 103.
The subtraction section 103 calculates a deviation Iref3 (=Iref2−Im) between the current command value Iref2 and a motor current Im of the motor 20 that is fed back, the deviation Iref3 is controlled by a current control section 104 such as a PI control (proportional and integral control) or the like, the controlled voltage control value E is inputted into a PWM control section 105 and the duty command value is calculated, and in accordance with a PWM-signal PS that the duty command value is calculated, the motor 20 is driven through an inverter 106. The motor current Im of the motor 20 is detected by a current detection circuit 120 within the inverter 106, and the detected motor current Im is inputted into the subtracting section 103 to feed back. In the case of vector-controlling the motor 20 by dq-axes, a resolver 21 as a rotation sensor is connected to the motor 20, and an angular speed calculating section 22 for calculating an angular speed co from a rotation angle θ is provided.
A bridge circuit that bridge-connects semiconductor switching elements (FETs) and the motor 20 is used in the inverter 106 that controls the motor current Im by means of the voltage control value E and drives the motor 20, and the motor current Im is controlled by performing ON/OFF control of the semiconductor switching elements in accordance with the duty command value of the PWM-signal determined based on the voltage control value E.
In the case that the motor 20 is a three-phase (U-phase, V-phase and W-phase) brushless DC motor, details of the PWM control section 105 and the inverter 106 become a configuration such as shown in FIG. 3. That is, the PWM control section 105 comprises a duty calculating section 105A that inputs each-phase carrier signals and calculates PWM-duty command values D1˜D6 of three phases (U-phase, V-phase and W-phase) in accordance with a predetermined expression based on the voltage control value E, and a gate driving section 105B that drives each gate of FET1˜FET6 by the PWM-duty command values D1˜D6 to turn ON/OFF. The inverter 106 comprises a three-phase bridge having upper/lower arms comprised of a U-phase upper-stage FET1 and a U-phase lower-stage FET4, upper/lower arms comprised of a V-phase upper-stage FET2 and a V-phase lower-stage FET5, and upper/lower arms comprised of a W-phase upper-stage FET3 and a W-phase lower-stage FET6, and drives the motor 20 by being turned ON/OFF with the PWM-duty command values D1˜D6. Further, electric power is supplied to the inverter 106 from the battery 13 through a power-source relay 14. The PWM-signal that determines ON/OFF timing of the switching elements for driving the multi-phase motor, is generated by comparing the carrier signal of a saw-tooth waveform or triangular waveform with the duty command value corresponding to a target current value in each phase of the multi-phase motor. Depending on the value of the carrier signal is equal to or more than the duty command value or the value of the carrier signal is less than the duty command value, the PWM-signal is determined as a high level (H) or a low level (L).
In such a configuration, although it is necessary to measure a drive current of the inverter 106 or the motor current of the motor 20, as one of request items of downsizing, weight saving and cost-cutting of the control unit 100, it is singularity of the current detection circuit 120. A one-shunt type current detection circuit is known as the singularity of a current detection circuit, and for example, the configuration of the one-shunt type current detection circuit 120 is shown in FIG. 4 (for example, Japanese Published Unexamined Patent Application No. 2009-131064 A). That is to say, a shunt resistor R1 is connected between the lower-stage arm of the FET bridge and ground (GND), a fall voltage that is caused by the shunt resistor R1 when a current flowed in the FET bridge is converted into a current value Ima by an operational amplifier (a differential amplification circuit) 121 and resistors R2˜R4, and further the current value Ima is A/D-converted at a predetermined timing by an A/D converting section 122 through a filter comprised of a resistor R6 and a capacitor C1, and then a current value Im that is a digital value is outputted. Moreover, a voltage “2.5V” being a reference voltage is connected to a positive input terminal of the operational amplifier 121 through a resistor R5.
FIG. 5 shows a wiring diagram of the battery 13, the inverter 106, the current detection circuit 120 and the motor 20 and also shows current routes (indicated by dashed lines) during a state that the U-phase upper-stage FET1 is turned ON (the U-phase lower-stage FET4 is turned OFF), the V-phase upper-stage FET2 is turned OFF (the V-phase lower-stage FET5 is turned ON), and the W-phase upper-stage FET3 is turned OFF (the W-phase lower-stage FET6 is turned ON). Further, FIG. 6 shows current routes (indicated by dashed lines) during a state that the U-phase upper-stage FET1 is turned ON (the U-phase lower-stage FET4 is turned OFF), the V-phase upper-stage FET2 is turned ON (the V-phase lower-stage FET5 is turned OFF), and the W-phase upper-stage FET3 is turned OFF (the W-phase lower-stage FET6 is turned ON). It is clear from these current routes of FIG. 5 and FIG. 6 that a total value of phases that the upper-stage FETs are turned ON, appears in the current detection circuit 120 as a detected current. That is, it is possible to detect a U-phase current in FIG. 5, and it is possible to detect the U-phase current and a V-phase current in FIG. 6. This is the same as in the case that the current detection circuit 120 is connected between the upper-stage arm of the inverter 106 and the power source (the battery 13). Moreover, in FIG. 5 and FIG. 6, the connection of the resolver 21 and the power-source relay 14 are omitted.
As a result, the current detection circuit 120 detects the current during a state that one phase is turned ON and during a state that two phases are turned ON, and by utilizing a characteristic that the sum of currents of three phases is “0”, it is possible to detect each-phase currents of three phases. Although a current IU of the U-phase is detected in the case of FIG. 5 and a total value of the current IU of the U-phase and a current IV of the V-phase is detected in the case of FIG. 6, since there is a relation of “IU+IV+IW=0” in the case of three phases, it is possible to detect the current IW of the W-phase as “IW=−(IU+IV)”.
However, in the inverter 106 that is configured by a single type current detection circuit 120 shown in FIG. 4, it is necessary to remove the influence of noise components such as rigging noises that occur due to a current flowing in the current detection circuit 120 just after each FET is turned ON and to detect an accurate current. Further, in the case that the interval of timing to turn the FET ON/OFF becomes very short between one phase and another phases, due to a matter that a necessary current for current detection does not flow in the FET, the existence of a dead time (a dead zone) and further a response delay of the circuit or the like, it becomes impossible to perform an accurate current measurement. In the case that the A/D converting section is used in the current detection circuit, the signal with the same magnitude has to be successively inputted for a fixed period (for example, 2 μs or more) so that the A/D conversion operation is carried out normally. This is because it is impossible for the A/D converting section to detect an accurate current value when a stable signal is not successively inputted.
Therefore, it is necessary to continue the state that one phase is turned ON and the state that two phases are turned ON only for a necessary time for the current detection. However, in the case that each-phase duty command values approximate each other, there is a problem that it is impossible to secure the necessary time for the current detection.
In the case that the time interval during switching of one phase and another phases is small, for example, by performing a correction to shift (arrangement-move) the phase of a predetermined phase, the time interval during switching of one phase and another phases becomes large, and it becomes possible to detect accurate current values of all phases of the multi-phase motor by means of a single type current detection circuit. However, as a result of performing the shift correction, when ON/OFF frequencies of the switching elements for driving the multi-phase motor are included within audio-frequency range, users hear the ON/OFF frequencies as noises (noisy sound) and the ON/OFF frequencies give an uncomfortable feeling to the users.
FIG. 7 shows a timing chart in the case that two phases are not detectable with respect to U-phase, V-phase and W-phase of three phases, one control period is 250 μs and includes five control periods of the PWM-signal of saw-tooth waveform of 50 μs period. FIG. 7 shows operations in the fourth and the fifth periods of the previous control period T1, and the first to the fifth periods of the present control period T2. The previous control period T1 shows a case that U-phase PWM-signal is duty 52%, V-phase PWM-signal is duty 47% and W-phase PWM-signal is duty 51%. Since a time interval between the V-phase being a duty minimum phase and the W-phase being a duty intermediate phase and a time interval between the W-phase being the duty intermediate phase and the U-phase being a duty maximum phase are 4% and 1% respectively, that is, since these two time intervals are short, if a phase shift is not performed, switching noises during the term do not go down, and an A/D conversion time for accurately detecting the current value cannot be secured. Therefore, the phase of the PWM-signal of the V-phase being the duty minimum phase is shifted to the left side (to advance phase) by 8%, and the phase of the PWM-signal of the U-phase being the duty maximum phase is shifted to the right side (to delay phase) by 11%. In this way, both the switching time interval between the V-phase and the W-phase and the switching time interval between the U-phase and the W-phase become large to 12%, and it is possible to detect accurate current values of the U-phase and the V-phase in each PWM-period.
The operations in the first to the fifth periods of the present control period T2 will be described. In the present control period T2, the U-phase PWM-signal decreases from duty 52% to duty 51%, the V-phase PWM-signal is duty 47% and there is no change in duty, and the W-phase PWM-signal increases from duty 51% to duty 52%. Therefore, the duty maximum phase changes from the U-phase to the W-phase, and the duty intermediate phase changes from the W-phase to the U-phase. Moreover, the duty minimum phase is the V-phase this time too. Since a time interval between the V-phase being the duty minimum phase and the U-phase being the duty intermediate phase and a time interval between the U-phase being the duty intermediate phase and the W-phase being the duty maximum phase are 4% and 1% respectively, that is, since these two time intervals are short, if the phase shift is not performed, switching noises during the period do not go down, and then A/D conversion time for accurately detecting the current value cannot be secured. Therefore, the phase of the PWM-signal of the V-phase being the duty minimum phase is shifted to the left side (to advance phase) by 8%, the phase of the PWM-signal of the W-phase being the duty maximum phase is shifted to the right side (to delay phase) by 11%, and the PWM-signal of the U-phase being the duty intermediate phase is not shifted.
In this way, in each of the five PWM-periods of the present control period T2, both the switching time interval between the U-phase and the V-phase and the switching time interval between the W-phase and the U-phase become large to 12%, and it is possible to detect accurate current values of the U-phase and the V-phase in each PWM-period.
Moreover, this example is a case that the U-phase changes from “shift” to “no shift”, the V-phase remains in “shift” and does not change in shift amount, and the W-phase changes from “no shift” to “shift”. Like this, when “shift”/“no shift” changes due to a change in the magnitude relation of each-phase duty ratios in the previous control period T1 and the present control period T2, at an ending time of the previous control period T1, i.e. at a starting time of the present control period T2, an instantaneous current fluctuation occurs as shown in a shunt waveform (a waveform of a voltage generated between both ends of a shunt resistor for current detection). In accordance with this drastic current fluctuation, there is a problem that noises from the motor that are based on a current ripple occur. Moreover, the shunt waveform of FIG. 7 shows the currents of the U-phase and −(the V-phase) in the previous control period T1 and also shows the currents of the W-phase and −(the V-phase) in the present control period T2.
As described above, in some cases, noises occur due to the influence of the current ripple associated with the drastic current fluctuation caused by a change in a shift state in respective control periods T1 and T2. The change in the shift state includes the following three cases.                (1) a change from “no shift” to “shift”        (2) a change from “shift” to “no shift”        (3) a change from “shift (shift amount A)” to “shift (shift amount B)” having a change of the shift amount (i.e. A≠B)        
As an apparatus or a method that solves such a problem, there is a multi-phase motor control apparatus disclosed in the publication of Japanese Patent No. 4884356 B2 (Patent Document 1). The multi-phase motor control apparatus of Patent Document 1 comprises a driving section comprising pairs of upper-stage arm switching elements and lower-stage arm switching element, and driving the multi-phase motor; a single type current detection circuit detecting current values of the multi-phase motor; a PWM-signal generating section generating plural each-phase PWM-signals within one control period based on the current value detected by the current detection circuit and a carrier signal; and a phase movement section moving the PWM-signal of a predetermined phase generated by the PWM-signal generating section by gradually changing a movement amount of the phase within one control period and outputting the PWM-signal which is moved to the driving section. Then, the phase movement section gradually increases a shift amount from zero in a control period of the present time in a case that a movement amount of the phase of the predetermined phase in a control period just before is zero and a movement amount of the phase of the predetermined phase in the control period of the present time is not zero. Further, the phase movement section gradually decreases the shift amount to zero in the control period of the present time in a case that the movement amount of the phase of the predetermined phase in the control period just before is not zero and the movement amount of the phase of the predetermined phase in the control period of the present time is not zero.